System and method for adaptive broadcast radar system

ABSTRACT

A method for obtaining target parameters within an adaptive broadcast radar system. In one embodiment, the method includes: coding information about a signal waveform generated by a transmitter having sub-apertures; receiving a signal at a receiver having sub-apertures corresponding to the sub-apertures of the transmitter, wherein the received signal correlates to the signal waveform; decoding information about the signal waveform from the received signal; and, determining a data quad from the decoded information, wherein the data quad includes degrees of freedom associated with the transmitter.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a divisional of U.S. patent application Ser. No.10/987,680 filed Nov. 12, 2004 now U.S. Pat. No. 7,053,821, and entitled“SYSTEM AND METHOD FOR ADAPTIVE BROADCAST RADAR SYSTEM”, which is acontinuation application of U.S. patent application Ser. No. 09/994,921(now U.S. Pat. No. 6,861,976) filed on Nov. 28, 2001, and entitled“SYSTEM AND METHOD FOR ADAPTIVE BROADCAST RADAR SYSTEM”, which claimsthe benefit of priority under 35 U.S.C. § 119 to U.S. Provisional PatentApplication No. 60/253,095 entitled “Adaptive Broadcast Radar”, filedNov. 28, 2000. Each of the foregoing is incorporated herein by referencein its entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a radar system and method, and moreparticularly, to a system and method for performing adaptive broadcastradar operations.

2. Discussion of the Related Art

Radar systems may be represented by a bistatic or multistatic radarsystem. A multistatic radar system has many receivers that are separatedfrom one or more transmitters. The radiated signal from a transmitterarrives at a receiver via two separate paths. One path may be a directpath from the transmitter to the receiver, and the other path may be atarget path that includes an indirect path from the transmitter to atarget to the receiver. Measurements may include a total path length, ortransit time, of the target path signal, the angle of arrival of thetarget path signal and the frequency of the direct and target pathsignals. A difference in frequency may be detected if the target is inmotion according to a doppler effect.

Knowledge of the transmitted signal is desirable at the receiver ifinformation is to be extracted from the target path signal. Thetransmitted frequency is desired to determine the doppler frequencyshift. A time or phase reference also is desired if the total scatteredpath length is to be determined. The frequency reference may be obtainedfrom the direct signal. The time reference also may be obtained from thedirect signal provided the distance between the transmitter and thereceiver is known.

Multistatic radar systems may be capable of determining the presence ofa target within the coverage of the radar, the location of the targetposition, and a velocity component, or doppler, relative to the radar.The process of locating the target position may include a measurement ofa distance and the angle of arrival. The measurement of distancerelative to the receiving site may desire both the angle of arrival atthe receiving site and the distance between transmitter and receiver. Ifthe direct signal is available, it may be used as a reference signal toextract the doppler frequency shift.

Known radar systems may transmit a signal beam in a specific directionto search for targets. Once a target has been detected, the beam may bedirected to follow the target. The receiver may receive scatteredsignals reflected off the target. By knowing the transmitter beamparameters, the receiver may perform operations to determine the targetparameters, as disclosed above.

Future airborne radar systems may operate in a difficult environmentwhere the detection of small and maneuverable targets may occur againsta strong clutter background and jamming operations. Directed beams ofenergy from transmitters may be susceptible to jamming countermeasuresand detection. Power aperture increases may not be effective to overcomethese limitations and countermeasures against radar detection. Thus,future systems may desire increase sensitivity without increasing powerrequirements. This condition may be applicable especially to radarsystems where the transmitter power is not controlled by the receivingparty.

Mobile radar systems often operate in the presence of jamminginterference and monostatic clutter that produced naturally by groundreflections. Difficulties may arise if both the transmitter and receiverare in motion, such as an airborne radar systems. When both thetransmitter and receiver of a radar system are in motion, the rank ofthe clutter covariance may be increased. An increased number of degreeof freedom in the receiver system may be needed to achieve a specifiedlevel of clutter suppression. Thus a transmitter or receiver in motionmay increase the clutter interference with a signal, or increase thecomplexity within the receiver in accounting for the increased degreesof freedom.

SUMMARY OF THE INVENTION

Accordingly, the present invention is directed to multistatic radarapplications and signal processing. Thus, a system and method foradaptive broadcast radar operations is disclosed herein.

According to a disclosed embodiment, a method for formatting receiveddata within an adaptive broadcast radar system having a transmittercomprising sub-apertures and a receiver comprising sub-apertures isdisclosed. The data is received at the receiver. The method includesproviding an estimate for a delay of scattered signal components withinthe received data. The method also includes generating an index for theestimate. The index may include a transmitter element number and areceiver element number. The method also includes generating a data quadfor the index. The method also includes estimating a measurementcovariance and a weight vector for the data quad. The data quad isreformatted with the measurement covariance and the weight vector.

According to another disclosed embodiment, a method for obtaining targetparameters within an adaptive broadcast radar system is disclosed. Themethod includes coding information about a signal waveform generated bya transmitter having sub-apertures. The method also includes receiving areceived signal at a receiver having sub-apertures corresponding to thesub-apertures of the transmitter. The signal correlates to the signalwaveform. The method also includes decoding information about the signalwaveform from the received signal. The method also includes determininga data quad from the information. The data quad may include degrees offreedom associated with the transmitter.

According to another disclosed embodiment, a method for generating asensor signal for a received signal within an adaptive broadcast radarsystem. The method includes defining a clutter component for thereceived signal at a receiver. The clutter component comprises a directpath signal and a scattered signal. The method also includes defining achannel transfer function. The method also includes generating a sampledversion of the received signal according to the channel transferfunction at a sample time. The method also includes determining a batchof data from the sampled version for a sub-aperture of the receiver atthe sample time. The method also includes indexing the batch of datainto the sensor signal model.

According to another embodiment, a method for transmitting a signalwaveform from a transmitter within an adaptive broadcast radar system isdisclosed. The transmitter comprises at least one sub-aperture. Themethod includes generating the signal waveform at the at least onesub-aperture. The method also includes coding the signal waveform at theat least one sub-aperture. The signal waveform is coded with thetransmitter data. The method also includes phase shifting the signalwaveform at the at least one sub-aperture. The method also includestransmitting the coded signal waveform from an array element coupled tothe sub-aperture according to the phase shifting.

According to another embodiment a method for performing radar operationwithin an adaptive broadcast radar system is disclosed. The radar systemincludes a transmitter having a first plurality of sub-apertures and areceiver having a second plurality of sub-apertures. The method includesencoding data on a signal waveform at a transmitter. The data includes anumber for said sub-apertures of the transmitter and degrees of freedomfor the transmitter. The method also includes continuously transmittingthe signal waveform. The method also includes determining a delay valueand a doppler value for received signals at said receiver. The receivedsignals comprise direct and scattered signals of the signal waveform.The method also includes regenerating a transmit signal beam correlatingto the signal waveform from the data, the delay value, and the dopplervalue.

According to another embodiment, an adaptive broadcast radar system isdisclosed. The radar system includes a transmitter comprising a firstplurality of sub-apertures. Each sub-aperture codes a signal waveformwith data. The radar system also includes a receiver comprising a secondplurality of sub-apertures coupled to a signal processor, wherein thesignal processor generates a transmit beam signal according to the datawithin each signal waveform.

Additional features and advantages of the invention will be set forth inthe description which follows, and in part will be apparent from thedescription, or maybe learned by practice of the invention. Theobjectives and other advantages of the invention will be realized andattained by the structure particularly pointed out in the writtendescription and claims hereof as well as the appended drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which is included to provide furtherunderstanding of the invention and is incorporated in and constitutes apart of this specification, illustrates embodiments of the presentinvention and together with the description serves to explain theprinciples of the invention. In the drawings:

FIG. 1 illustrates a block diagram of an adaptive broadcast radar systemfor detecting and tracking a target in accordance with an embodiment ofthe present invention;

FIG. 2 illustrates a transmitter within an adaptive broadcast radarsystem in accordance with an embodiment of the present invention;

FIG. 3 illustrates a receiver within an adaptive broadcast radar systemin accordance with an embodiment of the present invention;

FIG. 4 illustrates a channel demultiplexer in accordance with anembodiment of the present invention;

FIG. 5 illustrates a signal processor in accordance with an embodimentof the present invention;

FIG. 6 illustrates a flowchart for formatting received data inaccordance with an embodiment of the present invention;

FIG. 7 illustrates a flowchart for generating a sensor signal model inaccordance with an embodiment of the present invention; and

FIG. 8 illustrates a flowchart for performing radar operations within anadaptive broadcast radar system in accordance with an embodiment of thepresent invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Reference will now be made in detail to the preferred embodiments of thepresent invention, examples of which are illustrated in the accompanyingdrawings.

FIG. 1 depicts a block diagram of an adaptive broadcast radar system fordetecting and tracking a target in accordance with an embodiment of thepresent invention. Radar detection system 10 includes a receiving system100 to track one or more targets of interest 150 by exploiting signalsfrom a plurality of transmitters 110, 112, and 114. Radar detectionsystem 10 also may be known as an adaptive broadcast radar system.

Receiving system 100 represents a family of multi-static wide areatarget surveillance sensors. Receiving system 100 system exploitcontinuous wave (“CW”) electromagnetic energy. Preferably, receivingsystem 100 may receive transmissions from a plurality of transmitters110, 112, and 114. Preferred embodiments of the transmitters for use inan adaptive broadcast radar system are disclosed below in greaterdetail. Transmitters 110, 112, and 114, however, may include any device,system or means to transmit uncontrolled signals.

Transmitters 110, 112, and 114 may transmit wideband electromagneticenergy transmissions in all directions. Some of these transmissions arereflected by one or more targets of interest 150 and received by PCLsystem 100. For example, reflected transmission 130 may be reflected bytarget 150 and received by receiving system 100. Further, with regard totransmitter 114, reference transmission 140 may be received directly byreceiving system 100. Receiving system 100 may compare referencetransmission 140 and reflected transmission 130 to determine positionalinformation about one or more targets of interest 150. Referencetransmission 140 also may be known as a direct path signal. Reflectedtransmission 130 also may be known as a target path signal. Positionalinformation may include any information relating to a position of target150, including location, velocity, and acceleration from determining atime difference of arrival (“TDOA”), a frequency difference of arrival(“FDOA”) and an angle of arrival (“AOA”).

Receiving system 100 may comprise different components, includingreceiver 102 and processing unit 104. According to the disclosedembodiments, transmitter 111 may be a transmitter array, while receiver102 may be a receiver array. Transmitter 111 may include a plurality ofelements such that each element transmits an independent signal. Thesignals may comprise orthogonal or pseudo-orthogonal signals. Each ofthe plurality of elements for the transmitter and the receiver maycomprise a dipole with a back-plane. Transmitter 111 may be in motion,and not at a fixed position.

Receiver 102 may be a moving receiver array that includes a plurality ofelements such that each element is configured to receive a scatteredsignal. Further, receiver 102 is configured to receive a set ofinformation relating to the independent signals of each of the firstplurality of elements. The radar functions of the disclosed embodimentsmay be performed using information received by any receiver withsuitable receive equipment and knowledge of the transmitter waveformcodes. Receiver 102 may be in motion, and not at a fixed position.Further, there may be a plurality of receivers, with all the receiversin motion. The receivers may not be coupled together, or incommunication, so as to act independently of each other. For example,each receiver may be on moving vehicles within a certain area ofemphasis. The receivers within the area may receive the transmittedsignals continuously from the transmitters, such as transmitter 111.

Processing unit 104 is configured to receive information from receiver102 and to determine target 150 location based on the scattered signalsand the set of information relating to the independent signals of eachof the first plurality of elements. Processing unit 104, along withreceiver 102, may be an adaptive array used in conjunction with a set ofantennae coupled to receiving system 100 to provide a versatile form ofspatial filtering. Processing unit 104 may combine spatial samples of apropagating field with signals 140 with a variable set of weights. Theweights may be chosen to reject interfering signals and noise.Specifically, the spatial filtering capability of the array mayfacilitate cancellation of hostile jamming signals and suppression ofclutter.

Processing unit 104 may reformulate each of the independent signalsprovided by receiver 102. Accordingly, each processing unit 104 may formindependently all potential beams generated by transmitter 111, or anysubset of beams generated by transmitter 111. Because processing unit104 does not control transmitter 111, a single transmitter may beutilized by multiple receiver/processing unit combinations. Thus,signals from a single transmitter may be recreated at each receiver,independently of the transmitter and the other receivers.

Processing unit 104 may be located physically with receiver 102 suchthat one processing unit is collocated or integral with each receiver.Alternatively, other arrangements may be possible, such as remotelylocating processing unit 104 from receiver 102.

According to the disclosed embodiments, radar system 10 may provide themechanism to obtain radar parameters, such as ground moving targetindication (“GMTI”), air moving target indication (“AMTI”), andsynthetic aperture radar (“SAR”) imaging by forming simultaneoustransmitter and receiver beams. The gain and directivity of each of thesimultaneous transmitter and receiver beams may be comparable to knownsystems, such as phased array radar and bistatic radar technology. Usingradar system 10, the transmitter and receiver beamforming may becontrolled by a user within the field of view of the transmitter,provided the user has knowledge of the radar waveform codes. Thus, radarfunctions may be provided on-demand over wide geographical areas. Theradar transmitter may be shared by multiple users over a widegeographical area without the need for specific requirements to task thesource of illumination. For example, referring to FIG. 1, transmitter111 and receiver 102 are not coupled to each so as to exchange data.

The transmitter and receiver beams may be formed and adapted after theradar signals have been digitized by receiver 102 within receivingsystem 100. For AMTI and GMTI functions, displaced phase center aperture(“DCPA”) and space-time adaptive processing (“STAP”) algorithms may beapplied to transmitter and receiver degrees of freedom. STAP algorithmsmay help mitigate interference and clutter problems within receivingsystem 100. The availability of degrees of freedom that are physicallylocated at the transmitter may be used to motion compensateindependently the transmitter and receiver in air and space basedsystems. As a result, clutter may be suppressed in an airborne or spacebased system with as few as four degrees of freedom. The distribution ofdegrees of freedom between the transmitter and receiver may be used toextend other clutter suppression techniques, such as doppler nulling,from monostatic and bistatic systems. Doppler nulling may be defined aseliminating spatially and spectrally concentrated noise. For SARfunctions, users may form single transmitter beams that remain focusedon a user-specified center for spotlight imaging. Alternatively,transmitter beams may be formed to support scan mode imaging.

STAP performance may be dependent on scattered interference, availabledegrees of freedom, available processing power, and cost. STAPoperations may take advantage of all the information available to radarsystem 10 to cancel interference adaptively. A STAP enabled radar system10 may be able to dynamically respond to changes in the interferenceenvironment. Independent channels in space or time may be referred to asdegrees of freedom. A STAP enabled system collects information from theindependent channels and may use the information to compute the optimumweighting to accomplish the goal. STAP processing may minimize, orcancel, clutter and jamming while preserving the target signal in adesired direction. A reduction of minimum detectable velocity may bepossible using STAP.

The primary degrees of freedom are in space and time. Spatial degrees offreedom are provided by the outputs of the array elements of radarsystem 10. Time degrees of freedom are formed by delayed replicas of theoutputs from the array elements, or time taps. Other potential STAPdegrees of freedom may include beam outputs and Doppler filter outputs(for post-Doppler or Beamspace STAP). Processing requirements are alsoan important factor in the cost of STAP. Too many adaptive degrees offreedom may overwhelm typical radar system signal processors.Transmitter degrees of freedom also may depend on the number ofelements, or sub-apertures, associated with the transmitter array.

Thus, the disclosed embodiments may use “noise-like” waveforms for radarto extend the coverage area. Further, the disclosed embodiments maydistribute spatially the radar degrees of freedom to obtain an increasedlevel of clutter and interference suppression. Alternatively, thedisclosed embodiments may enable clutter suppression with a smallernumber of degrees of freedom than known systems. Clutter andinterference suppression may be achieved by using orthogonal andpseudo-orthogonal transmitted waveforms to provide a dual transmit andreceive aperture adaptivity. This feature may enable a reduction in therequired sensor system degrees of freedom.

According to the disclosed embodiments, an adaptive broadcast radarsystem is disclosed that reduces the degrees of freedom within thesystem and enables the spatial distribution of degrees of freedombetween a transmitter and a receiver. For example, the disclosedembodiments may be used to suppress clutter in an airborne orspace-based system with as few as four degrees of freedom. Similarly,these degrees of freedom may be distributed spatially between thetransmitter and the receiver according to the prevailing circumstances.A more robust transmitter may be implemented in conjunction with a lessrobust set of receivers. This configuration may be desired for thosesystems adapted to a surveillance environment.

Radar system 10 may share transmitter resources among a wide variety ofusers such that a priori tasking or the control of transmitter resourcesis not critical. Additional processing, intelligence, or tasking isavoided at the transmitter, other than operations to code independentwaveforms. Radar functions may be performed against objects, or targets,anywhere within the field-of-view of the independently coded sub-arrayswithin the transmitter. Where coding occurs at the element level, theradar functions may be performed against objects anywhere in the forwardhemisphere of the transmitter array by any user with suitable receivingequipment and knowledge of the transmitter waveform codes. Thetransmitted signal waveforms may be reconstructed at the receiever,thus, possibly eliminating the need for a directed transmitted beam.

The transmitter waveforms are regenerated at the receiver and each usermay form all potential beams independently, or any subset of potentialtransmitter beams. Thus, the transmitter waveforms are recreated at thereceiver without being coupled directly to the transmitter. Thetransmitter beam pattern may be adapted by a user for purposes thatinclude spotlight or scan mode SAR imaging, and interference/cluttersuppression. Adaptive processing may include motion compensation of thetransmitter. When combined with the motion compensation of the receiver,adaptive processing may provide bistatic displaced phase center aperture(“DPCA”) clutter suppression.

The transmitter antenna phase centers may be added to the degrees offreedom for STAP algorithms. Data for STAP processing is organized into3-dimensional arrays, where the 3-dimensional index identifies thetransmitter/receiver array element number, the relative delay anddoppler associated with a batch or coherent processing interval of data,and the time associated with the batch or coherent processing intervalof data. The three-dimensional array may be known as data cubes. Theinclusion of additional degrees of freedom associated with thetransmitter array transforms a data cube into a new data structuretermed a data quad. The STAP formulation enables a large class ofemerging STAP techniques and algorithms to be developed for monostaticradars to be directly reformulated for bistatic radar systems, such asradar system 10.

FIG. 2 depicts a block diagram of a transmitter for an adaptivebroadcast radar system in accordance with an embodiment of the presentinvention. Transmitter 200 may be used in the adaptive broadcast radarsystem disclosed above. Transmitter 200 includes sub-apertures 210, 220,and 230. Transmitter 200 may have N number of sub-apertures, and is notlimited to the number disclosed with reference to FIG. 2. Sub-apertures210, 220, and 230 include waveform generators 212, 222, and 232,respectively. Waveform generators 212, 222, and 232 produce independentpseudo-random phase samples. Clock 202 is coupled to wave formgenerators 212, 222, and 232.

In addition to waveform generator 212, sub-aperture 210 includes timeaperture 214 and amplifier 216. Sub-aperture 210 also includes phaseshifters 218 and sub-aperture weights 219. In addition to waveformgenerator 222, sub-aperture 220 includes time aperture 224 and amplifier226. Sub-aperture 220 also includes phase shifters 228 and sub-apertureweights 229. In addition to waveform generator 232, sub-aperture 230includes time aperture 234 and amplifier 236. Sub-aperture 232 alsoincludes phase shifters 238 and sub-aperture weights 239. Sub-aperture230 may be the Nth aperture within transmitter 200.

Transmitter 200 preferably is a phased array antenna with multiple phasecenters, and with independent signals transmitted on each element, orsub-aperture, of the array. The set of transmitted signals may beselected to orthogonal or pseudo-orthogonal. Orthogonal orpseudo-orthogonal signals may be generated using waveform coding, suchas Gold codes. The signal spectrum may be shared between transmitelements, or sub-apertures, using code division multiple access(“CDMA”). Alternatively, the transmit signals may be assigned toindependent frequency channels, and the spectrum shared using frequencydivision multiple access (“FDMA”). FDMA approaches also may includeguard bands within the signal spectrum to accommodate anticipateddoppler shifted clutter and targets. Thus, near orthogonality may becompleted. Polarization state also may be used to define orthogonaltransmit channels.

Thus, sub-aperture 210 tasks waveform generator 212 to generate awaveform from independent pseudo-random phase samples. The waveform ispassed to time aperture 214. Time apertures may create a train of pulsesthat are coded differently for each sub-aperture. Alternatively, timeaperture 214 may be bypassed to provide continuous wave (“CW”) codedsignals. For example, pseudo-random phase coding may occur where thelength of the sequence exceeds the expected coherent correlationintervals. The waveform or train of pulses is received by amplifier 216and passed to a bank of phase shifters 218 and sub-aperture weights 219.Phase shifters 218 and sub-aperture weights 219 may be set independentlyfor each sub-aperture. Phase shifters 218 and sub-aperture 219 may bedesigned to generate a fixed beam in a direction relative to theboresight of transmitter 200. Further, the waveform is coded with theinformation to generate the data quads at the receiver, as disclosed ingreater detail below. The waveform may be forwarded to antennas 280 fortransmission. Sub-apertures 220 and 230 may operate in a similar manner,such that different waveforms are transmitted by antennas 280.

The formation of sub-apertures allows the number of degrees of freedomto be limited, and, thus, reduce system complexity. A correspondingreduction in coverage may occur where the coverage area is defined bythe sub-aperture beam patterns. An alternative approach to reduce systemcomplexity may involve operations at lower frequencies. The use of lowerfrequencies may reduce the number of transmitter elements desired forfull coverage in the forward hemisphere of transmitter 200.

FIG. 3 depicts a block diagram of a receiver for an adaptive broadcastradar in accordance with an embodiment of the present invention.Receiver 300 may receive transmitted signals directly or indirectly fromtransmitter 200 with antennas 380. Receiver 300 includes a clock 302 andsub-apertures 310, 320, and 330. Receiver 300 may have N number ofsub-apertures, and is not limited to the number disclosed with referenceto FIG. 3. Like transmitter 200, receiver 300 employs banks of phaseshifters and sub-aperture weights within sub-apertures 310, 320, and 330to modify the received signals from antennas 380. The formation of thesub-apertures may limit the number of degrees of freedom to reducesystem complexity.

Sub-apertures 310, 320, and 330 also include receivers 316, 326, and336, respectively. Receivers 316, 326, and 336 may be low-noise, highdynamic range receivers. In receivers 316, 326 and 336, sub-apertureformation may be used to reduce cost and complexity of the radar systemif the adaptive broadcast radar transmitter, such as transmitter 200, isutilized for surveillance or reconnaissance in a restricted area. Tosupport operations that desire wide area coverage, the radar system mayoperate at a lower frequency to reduce system complexity without acorresponding reduction in coverage. Receivers 316, 326, and 336 arecoupled to adaptive broadcast radar transmitter channel demultiplexers350, 360, and 370, respectively. Preferably, demultiplexer is dedicatedto each sub-aperture.

The sampled output of the k^(th) sub-aperture in the receiver system maybe denoted V_(Rx) (k, t_(n)). Motion compensation functions 318, 328,and 338 may remove time dependent phase delays between the transmitterand receiver system. Motion compensation may be performed for eachreceiver sub-aperture independently. Because the received signal is acomposite of transmitted signals, a single point on the transmitter,such as transmitter 200, may be motion compensated. The transmitter maybe known as the j₀ ^(th) transmitter sub-aperture. To simplify thederivations, a scaling also may be included in motion compensationfunctions 318, 328, and 338. The scale factor may be the inverse of thetransmitted signal strength. The signal strength may be proportional tothe square-root of the transmitter power delivered to the j₀ ^(th)transmitter sub-aperture. The motion compensated signal may be given by:

${X_{k}(t)} \equiv {\frac{V_{RX}( {k,t} )}{V_{Tx}(t)}{\mathbb{e}}^{{- 2}{\pi\mathbb{i}}\;{f{({t - {\tau{({{{\overset{\_}{x}}_{{Tx}:{j0}}{(t)}} - {{\overset{\_}{x}}_{{RX}:k}{(t)}}})}}})}}}}$where f is the center frequency of the transmitted signal, and, τ( X_(Tx:j0)(t)− x _(Rx:k)(t)) is the signal propagation delay from the j₀^(th) transmitter sub-aperture to the k^(th) receiver sub-aperture. Forexample, transmitter sub-aperture 210 may send a signal to receiversub-aperture 310 that is motion compensated. The above algorithmdiscloses the motion compensation operation performed by motioncompensation function 318.

Receiver 300 also includes signal processor 390. Signal processor 390may be coupled to the sub-apertures of receiver 300, such assub-apertures 310, 320, and 330. Signal processor 390 may correlate toprocessing unit 104 depicted in FIG. 1. Signal processor 390 forms thedata quads for STAP operations within the adaptive broadcast radarsystem. Signal processor 390 is disclosed in greater detail below. Usingthe respective demultiplexers and signal processor 390, the transmittedsignal waveform may be regenerated from the information encoded onto thewaveform at transmitter 200. Receiver 300, as disclosed in greaterdetail below, is capable of recreating the transmit signal beam from thenumerical data encoded on the waveform, such as degrees of freedom oftransmitter 200 without being directly coupled to transmitter 200.

FIG. 4 depicts a block diagram of an adaptive broadcast radar channeldemultiplexer in accordance with an embodiment of the present invention.Demultiplexer 400 correlates to demultiplexers 350, 360, and 370depicted in FIG. 3. Demultiplexers 350, 360, and 370, however, are notlimited to the embodiments disclosed by FIG. 4. Demultiplexer 400includes a bank of waveform generators 402, 404, and 406. Waveformgenerators 402, 404, and 406 are time-synchronized with the transmitter,such as transmitter 200, and may generate replicas of each of the Ntransmitted signals, where N is the number of transmitter sub-apertures.Thus, demultiplexer 400 may have N waveform generators correlating to Nsub-apertures of transmitter 200. Waveform generators 402, 404, and 406are coupled to clock 302.

Waveform compensation may be initialized and/or updated once percoherent processing interval. The coherent processing interval is chosensuch that the number of signal samples is greater than or equal to J, #or the number of transmitter sub-apertures, times M, or the number ofdelay values desired to cover the ground clutter grid. A waveformcompensation filter computation function 408 may generate and format anN_(t)×(J·M) array of delayed reference signals, where N_(t) may be thenumber of samples in a coherent processing interval. The referencesignal data, s_(j) (t_(n)−τ_(m)), associated with the m^(th) delay forthe j^(th) transmitter sub-aperture is mapped into the q^(th) column,where q(j,μ)=μ+(j−1)·M. The inverse map of the generalized index, q,into the sub-aperture index, j, and the delay index, μ, may be given by:

μ ≡ mod(q, M) $j \equiv {{floor}( \frac{q}{M} )}$

For the coherent processing interval starting with the time samplet_(n0), the array of delay-compensated reference data may be given by:Σn,q(n ₀)≡s _(j(q))(t _(n)−τ_(μ(q)))

The term also may be written in terms of {tilde over(S)}_(j,n)≡S_(j)(t_(n)). Because t_(n)−τ_(μ(q))=t_(n−μ(q)) andS_(j)(t_(n)−τ_(μ))={tilde over (S)}j,n−m, the array of delay-compensatedreference may be given by:

${\sum({n0})} \equiv \begin{pmatrix}{\overset{\sim}{S}}_{0,{n\; 0}} & {\overset{\sim}{S}}_{0,{{n\; 0} - 1}} & \cdots & {\overset{\sim}{S}}_{0,{{n\; 0} - M + 1}} & {\overset{\sim}{S}}_{1,{n0}} & {\overset{\sim}{S}}_{1,{{n\; 0} - 1}} & \cdots & {\overset{\sim}{S}}_{0,{{n\; 0} - M + 1}} & \cdots & {\overset{\sim}{S}}_{J,{n\; 0`}} & {\overset{\sim}{S}J_{,{{n\; 0} - 1}}} & \cdots & {\overset{\sim}{S}J_{,{{n\; 0} - M + 1}}} \\{\overset{\sim}{S}}_{0,{{n\; 0} - 1}} & {\overset{\sim}{S}}_{0,{{n\; 0} - 2}} & \cdots & {\overset{\sim}{S}}_{0,{{n\; 0} - M}} & {\overset{\sim}{S}}_{1,{{n\; 0} - 1}} & {\overset{\sim}{S}}_{1,{n\; 0}} & \cdots & {\overset{\sim}{S}}_{0,{{n\; 0} - M}} & \cdots & {\overset{\sim}{S}J_{,{{n\; 0} - 1}}} & {\overset{\sim}{S}J_{,{n\; 0}}} & \cdots & {\overset{\sim}{S}J_{,{{n\; 0} - M}}} \\\cdots & \cdots & \cdots & \cdots & \cdots & \cdots & \cdots & \cdots & \cdots & \cdots & \cdots & \cdots & \cdots \\{{\overset{\sim}{S}}_{0,{{n\; 0} + {Nt}}} - 1} & {\overset{\sim}{S}}_{0,{{n\; 0} + {Nt} - 2}} & \cdots & {\overset{\sim}{S}}_{0,{{n\; 0} + {Nt} - M}} & {\overset{\sim}{S}}_{1,{{n\; 0} + {Nt} - 1}} & {\overset{\sim}{S}}_{1,{{n\; 0} + {Nt}}} & \cdots & {\overset{\sim}{S}}_{0,{{n\; 0} + {Nt} - M}} & \cdots & {\overset{\sim}{S}J_{,{{n\; 0} + {Nt} - 1}}} & {\overset{\sim}{S}J_{,{{n\; 0} + {Nt} - 2}}} & \cdots & {\overset{\sim}{S}J_{,{{n\; 0} + {Nt} - M}}}\end{pmatrix}$

If N_(t) is selected such that N_(t)=J·M, summation Σ may be a squarematrix, and, for pseudo-random phase code signals, it may be shown to beinvertible. Accordingly, waveform compensation filter computationfunction 408 may be Σ⁻¹. Thus, when N_(t)>J·M, the pseudo-inverse may beused and waveform compensation filter 408 may be given by:(Σ^(*T)Σ)⁻¹Σ^(*T)

The output of waveform compensation filter 408, also known as thechannel transfer function, is given by H=W^(T)Y.

When the channel transfer functions H include both delayed anddoppler-shifted signal components, summation Σ may be replaced by anarray of reference signals that are compensated for both delay, τ_(μ),and doppler shift, f_(v). A generalized index q defines the column fordata associated with the j^(th) sub-aperture, the μ^(th) delay andν^(th) doppler andq(j,μ,v)≡v+[μ+(j−1)·M]·N

The inverse map, of the generalized index, q, into the sub-apertureindex, j, and delay index, μ, may be given

v ≡ mod(q − 1, N) + 1$\mu \equiv {{{mod}( {{{floor}( \frac{q - 1}{N} )},M} )} + 1}$$j \equiv {{{floor}( \frac{q - 1}{M \cdot N} )} + 1}$

Then, the array of compensated reference signals may be given by:

${\sum n},{{q( n_{0} )} \equiv {{\mathbb{e}}^{2{\pi\mathbb{i}}\;{{fv}{(q)}}{({{tn} - \tau_{\mu{(q)}}})}}{s_{j{(q)}}( {{tn} - \tau_{\mu{(q)}} - {\frac{\lambda\; f_{v{(q)}}}{c_{light}}( {t_{n} - \tau_{\mu{(q)}}} )}} )}}}$where nε[n₀, n₀+N_(t)−1] and qε[1, J·M·N−1].

For the k^(th) receiver system sub-aperture, H is a vector of lengthJ·M·N. Vector H may be reformatted into a J×(M×N) array where the j^(th)element discloses the dependence of the channel transfer function ontransmitter sub-array degrees of freedom and (μ,v) discloses the delayand doppler dependence.

Thus, according to the disclosed embodiments, demultiplexer 400 receivesthe motion compensated signals Y_(n) at input data format function 412.Input data format function 412 also receives synchronization data fromclock 302. Waveform generators 402, 404, and 406 also are synchronizedby clock 302 to generate replicas of the transmitted signals. Thesereplica signals are passed to waveform compensation filter function 408,also synchronized with clock 302. Waveform compensation filter function408 outputs compensated reference signals to channel transfer function410, which outputs channel transfer functions 490.

Channel transfer functions 490 contain delay, doppler, and otherinformation about the received signal data. Channel transfer functions490 may be specific to received signal data from each sub-aperture ofreceiver 300.

FIG. 5 depicts a block diagram of a signal processor in accordance withan embodiment of the present invention. Signal processor 500 enables theformation of data quads for STAP applications. Because of thesimultaneous transmitter and receiver beams, the steering vector, G, isa quad that discloses the desired sensor response for both transmitter200 and receiver 300. STAP may be a framework rather than a specificalgorithm. The definition of the steering vector and the definition ofthe approach to modeling and estimating the measurement covariance aredesired to transform the STAP framework into a specific algorithm.

The STAP covariance may be modeled as a diagonal matrix, independentlyof delay and time to provide simultaneous fixed transmitter and receiverbeamforming. Weight vectors are proportional to the steering vector andprovide the amplitude and phase adjustments desired to steer thetransmitter and receiver beams in a particular direction. Diverse typesof fully adaptive and partially adaptive STAP algorithms designed toadapt both transmitter and receiver antenna patterns may be specified interms of the weight vectors. For example, bistatic DPCA may beimplemented using steering vector components may be defined by:

$G = \begin{Bmatrix}g_{i} & \otimes & g_{k} \\{{- 2}g_{i}} & \otimes & g_{k} \\g_{i} & \otimes & g_{k}\end{Bmatrix}$

Thus, signal processor 500 may operate as follows. Channel transferfunctions 490 are received from demultiplexer 400 at channel transferdata function 512. Channel transfer data function 512 formats thechannel transfer functions 490 into channel transfer components, orH={h_(j,k)(t_(n m))}. Steering vector computation function 510 receivescoherent processing interval parameters from clock 302 and computessteering vector components, or G={g_(j,k)(t_(n m))}. STAP data quadreformat functions 516 and 514 then reformat the channel transfercomponents and steering vector components, respectively.

STAP covariance function 518 receives the channel transfer component Hand computes, or estimates, a measurement covariance. The STAPcovariance may be depicted as a diagonal matrix, and independent ofdelay and time to provide simultaneous fixed transmitter and receiverbeamforming. STAP weight vector function 520 receives the measurementcovariance and the steering vector component, G, to compute the weightvector, or W=R⁻¹G. Weight vectors may be proportional to the steeringvector, G. Weight vectors may provide the amplitude and phaseadjustments desired to steer the transmitter and receiver beams in aparticular direction. This feature allows diverse types of fullyadaptive and partially adaptive STAP algorithms for both transmitter andreceiver antenna patterns to be specified in terms of the weightvectors.

The weight vectors are received at filtered data function 522, alongwith channel transfer components, or H. The channel data within thechannel transfer components may be reformatted and the data quadscomputed by applying the weight vectors to the channel transfercomponents. Thus, data quad 530 is output to other system components forfurther processing. Data quad 530 may be defined in four-dimensions byindices determined by filtered data function 522.

FIG. 6 depicts a flowchart for formatting data within the receivedsignal in accordance with an embodiment of the present invention. Thedata is formated to form the data quads disclosed above. The signal maybe received at receiver 300, and the following disclosed operations maybe executed within receiver 300, demultiplexer 400, and signal processor500. Step 602 executes by compensating the receiver data from thesignal. Specifically, the receiver data is motion compensated byprocessing the data associated with each element, or sub-aperture, ofthe receive array of receiver 300. Step 604 executes by removing thedoppler shift from the receiver data. Step 604 may be executed inconjunction with step 602. The doppler shift may exist due to therelative motion of receiver 300 and transmitter 200. The doppler shiftremoval may be given byX_(k)(t)

X ₀ _(k) (t)e ^(−2πif) _(Tx−Rx) ^(t)

Step 606 executes by estimating the direct path component, ortransmitter signal, of the received signal. The total transmitted signalmay be modeled parametrically and the model parameters estimated. Thetransmitter signal may be modeled on an element-by-element basis and themodel may include parameters to describe the bearing of receiver 300relative to the boresight of transmitter 200, and the magnitude of thetransmitted signal power at the element of receiver 300. Modelparameters may be estimated using known adaptive techniques to minimizereceived power. Step 608 executes by cancelling the direct pathcomponent, or transmitter signal from the received signal. Specifically,estimates of the direct path signal, as disclosed above, may besubtracted from the received data, given by

${Y_{k}(t)} = {{X_{k}(t)} - {a_{k}{\sum\limits_{j = 1}^{j}{\{ {\mathbb{e}}^{2{{\pi\mathbb{i}}{({j - 1 - \frac{j - 1}{2}})}}\delta_{L}{\sin({(\varphi_{k})}}} \} S_{j}}}}}$

Step 610 executes by segmenting the received data into the coherentprocessing intervals. Coherent processing intervals also may be known as“dwells.” Step 612 executes by providing estimates of delay ordelay/doppler of the scattered signal components from the segmenteddata. The segmented data is coherently processed to provide theestimates. The estimates may be given by

$( ( \chi_{n} )_{m} )_{j,k} = {{\int_{T_{n}}{{{\overset{\_}{S}}_{j}( {t - \tau_{m}} )}{Y_{k}(t)}\ {\mathbb{d}{t( ( \chi_{n} )_{m,v} )}_{j,k}}}} = {\int_{T_{n}}{{{\overset{\_}{S}}_{j}( {t - \tau_{m}} )}{\mathbb{e}}^{{- 2}{{\pi\mathbb{i}}{({v - 1})}}\delta\; f_{dop}}{Y_{k}(t)}\ {\mathbb{d}t}}}}$

Step 614 executes by generating indices for the estimates determinedabove. The indices, j and k, may define the appropriate transmitter andreceiver element numbers, respectively. The indexing scheme is generalenough to accommodate two-dimensional arrays. Further, in the case ofthe second estimate equation disclosed above, the independent delay anddoppler indices may be combined into a single index, or m′=m+M_(τ)(n−1),where M_(τ) is the total number of delay measurements in the measurementdomain. Step 616 executes by generating the data quads according theindices. The indices, (j, k, m, n) may define points in thefour-dimensional data quads. Standard index mapping may be used totransform the dimensional data vector into a one-dimensional vector,X_(v). Step 618 executes by estimating the measurement covariance, R,for the vector disclosed above. Standard STAP techniques may be appliedto estimate the measurement covariance, or R_(vv′)=(x_(v)*x_(v′)). Step620 executes by computing the STAP weight vectors. The STAP weightvectors, W, may be computed in terms of the covariance and the steeringvector, G, or W=R⁻¹G. As disclosed with reference to FIG. 5, the weightvectors provide the amplitude and phase adjustments desired to steerbeams in a particular direction.

Step 622 executes by reformatting the data quad, and its information,with the weight vector. The data quad includes information on thetransmitter/receiver element number, the delay and/or doppler associatedwith the batch of data for the coherent processing interval, the timefor the coherent processing interval, and the degrees of freedom. Theinformation in the data quad describes the transmitted signal such thatit may be regenerated by receiver 300. Step 624 executes by forwardingthe data quad information to additional processing components.

The following examples may be applied to bistatic radar. First, anextension of DPCA may be considered. The extension may be designed toprovide adaptation in the transmit and receive arrays.

The equation Let

$g_{Tx} = \{ {\mathbb{e}}^{2{{\pi\mathbb{i}}{({k\frac{K + 1}{2}})}}\delta\;{\sin{(\varphi_{Tx})}}} \}$may represent the transmit aperture weights designed to provide a fixedbeam in direction, φ_(Tx), relative to the boresight of transmitter 200.In addition, the equation let

$g_{Rx} = \{ {\mathbb{e}}^{2\pi\;{i{({k\frac{K + 1}{2}})}}\delta\;{\sin{({\varphi\mspace{11mu}{Rx}})}}} \}$may define the desired weights for the receive system as depicted withreference to FIG. 3. G may be formed as follows:

$G = {\begin{Bmatrix}{g_{Tx} \otimes g_{Rx}} \\{{- 2}{g_{Tx} \otimes g_{Rx}}} \\{g_{Tx} \otimes g_{Rx}}\end{Bmatrix}.}$G may correspond to a three-pulse DPCA. Second, G may be defined as thesteering vector for a target doppler, f_(d), or

$G = \begin{Bmatrix}{{g_{Tx} \otimes g_{Rx}}\;{\mathbb{e}}^{{- 2}\;\pi\;{\mathbb{i}}\; f_{d}T}} \\{{- g_{Tx}} \otimes g_{Rx}} \\{{g_{Tx} \otimes g_{Rx}}\;{\mathbb{e}}^{2\;\pi\;{\mathbb{i}}\; f_{d}T}}\end{Bmatrix}$

FIG. 7 depicts a flowchart for generating a sensor signal model inaccordance with an embodiment of the present invention. A sensor signalmodel may facilitate removing the pseudo-random baseband modulation fromthe received signal. The sensor signal model may disclose the receivedsignal in terms of the clutter and target environment, and is desired tointerpret effectively the sensor output. Step 702 executes by defining aclutter component for the sensor signal model. According to thedisclosed embodiments,

${X_{p}( {k,t} )} = {{\sum\limits_{j = 0}^{J - 1}\{ {{{A_{{0:j},k}( {{\overset{\_}{x}}_{c},{\overset{\_}{x}}_{Tx},{\overset{\_}{x}}_{Rx}} )}\;{\mathbb{e}}^{{\mathbb{i}}\;{\psi_{jk}{(t)}}}} + {\int_{A_{clutter}}^{\;}{{A_{{c:j},k}( {{\overset{\_}{x}}_{c},{\overset{\_}{x}}_{Tx},{\overset{\_}{x}}_{Rx}} )}\;{\mathbb{e}}^{{- {\mathbb{i}}}\;{\phi_{c:{jk}}({{\overset{\_}{x}}_{c},t})}}{\mathbb{e}}^{{\mathbb{i}}\;{\psi_{jk}{({t - {\Delta\;{\tau_{c:{jk}}{({{\overset{\_}{x}}_{c},t})}}}})}}}{\mathbb{d}^{2}{\overset{\_}{x}}_{c}}}}} \}} + {v(t)}}$

where Φ^(j) is the phase of the baseband signal transmitted though thejth sub-aperture. Ψ^(j) _(k) may be equal to a delayed version of Φ^(j)where the delay may be about equal to the direct path delay from thereference transmitter sub-aperture, j₀, to the kth receiversub-aperture, orΨ_(k)(t)≡Φ^(j)(t−τ( x _(Tx:j) ₀ (t)−i x _(Rx:k)(t)))Δτ_(c:jk) (x_(c),t) may be the delay from transmitter 200 to clutterpatch to receiver 300 relative to the direct path delay as disclosedabove. In particular, the delay may be given byΔτ_(c:jk)(x _(c) ,t)≡τ(R _(ic)(t))+τ(R _(ck)(t))−τ(R _(0:j) ₀ _(k)(t))φ_(c:jk) (x_(c),t) may be the delay from transmitter 200 to clutterpatch to receiver 300 relative to the direct path phase delay. The RFphase delay may be given byφ_(c jk)( x _(c),t)≡2πf₀Δτ_(c:jk)( x _(c),t)A_(0:j,k) may be the relative strength of the direct path signalrelative to the transmitted signal strength. G_(Tx:jk) may denote thesub-aperture gain of the j^(th) transmitter sub-aperture in thedirection of the k^(th) receiver sub-aperture. G_(Rx:jk) may denote thesub-aperture gain of the k^(th) receiver sub-aperture in the directionof the j^(th) transmitter sub-aperture. L may denote the path loss forthe direct path signal component. The relative strength of the directpath signal relative to the transmitted signal strength may be given byA_(0;j,k)( x _(Tx), x _(Rx))≡└G_(Tx:jk)G_(Rx:jk)L(R_(0:jk))┘A_(c:j,k) may be the relative strength of the scattered signal where x_(c) may denote the location of the clutter patch, ê_(jc) and ê_(kc) maybe unit vectors that point from transmitter 200 to the clutter patch andfrom the clutter patch to receiver 300. σ⁰( x _(c)|ê_(jc),ê_(ck)) may bethe relative bistatic reflectivity of the ground patch at x _(c). Thus,the relative strength of the scattered signal may be given byA_(c:j,k)( x _(c), x _(Tx), x_(Rx))≡└G_(Tx)(ê_(jc))G_(Rx)(ê_(ck))L(R_(jc))L(R_(ck))σ⁰( x_(c)|ê_(jc),ê_(ck))In addition v(t) may denote the noise process.

Step 704 executes by generating integral of the received signal. Thereceived signal may be re-expressed in terms of a continuous integralover the delay measurement, τ, or

${X_{p}( {k,t} )} = {{\sum\limits_{j = 0}^{J - 1}\{ {{{H_{{0:j},k}(t)}\;{V_{jk}(t)}} + {\int_{\Delta\;\tau}^{\;}{{H_{{c:j},k}( {\tau^{\prime},t} )}{V_{jk}( {t - \tau^{\prime}} )}\;{\mathbb{d}\tau^{\prime}}}}} \}} + {v(t)}}$where V_(jk)(t)≡e^(iψ) ^(j) ^(k(t)).

Step 706 executes by defining the bistatic channel transfer function(“BCTF”). H_(c:j,k) (τ,t) may be termed the BCTF and may be given by anintegral over a constant delay strip, or

${H_{{c:j},k}( {\tau,t} )} \equiv {\int_{\lbrack{{\Delta\;{\tau_{c}{({\overset{\_}{x}}_{c})}}} = \tau}\rbrack}^{\;}{{\mathbb{e}}^{{- {\mathbb{i}}}\;{\varphi_{c:{jk}}{({{\overset{\_}{x}}_{c},t})}}}{A_{{c:j},k}( {{\overset{\_}{x}}_{c},{\overset{\_}{x}}_{Tx},{\overset{\_}{x}}_{Rx}} )}\;{\mathbb{d}^{2}{\overset{\_}{x}}_{c}}}}$

Step 708 executes by linearizing the phase delay of the BCTF.Linearization of the phase delay in the BCTF may demonstrate thedependence on doppler and the bearing of transmitter 200 and receiver300 to the clutter patch, orφ_(c:jk)( x _(c) ,t)=φ₀ +K ^(T) ·D _(p)where

$D_{p} = \begin{bmatrix}{{\,^{d}{Tx}}\;\lfloor {{\sin( {\,^{\varphi}{Tx\_ c}} )} + {\sin( {\,^{\varphi}{Tx\_ Rx}} )}} \rfloor} \\{{\,^{d}{Rx}}\lbrack {{\sin( {{\,^{\phi}{c\_ Rx}} + {\,^{\eta}{Rx}}} )} - {\sin( {{\,^{\phi}{Tx\_ Rx}} + {\,^{\eta}{Rx}}} )}} \rbrack} \\\begin{matrix}{{{{\,^{V}{Tx}}\lbrack {{\sin( {\,^{\phi}{Tx\_ c}} )} + {\sin( {\,^{\phi}{Tx\_ Rx}} )}} \rbrack}\;\delta\; t_{nyquist}} +} \\{{{\,^{V}{Rx}}\lbrack {{\sin( {{\,^{\phi}c} + {Rx}} )} - {\sin( {\,^{\phi}{Tx\_ Rx}} )}} \rbrack}\;\delta\; t_{nyquist}}\end{matrix}\end{bmatrix}$

Step 710 executes by absorbing the constant phase term into the relativestrength of the scattered signal, or A_(c:j.k). Thus, the BTCF may begiven byH_(c:j,k)(τ,t)≡∫e^(−iK) _(T·) ^(Dp)A_(c:j,k( x) _(c) _(, x) _(Tx) _(, x)_(Rx) _()d) ²⁻ _(x) _(c) .[Δτ_(c)( x _(c))=τ]

Step 712 executes by defining the channel transfer function. The signalmay be sampled at the in-time at the Nyquist rate and in delay at a rateconsistent with the delay resolution, δτ. Step 714 executes bygenerating a sampled version of the received signal. The sampled versionof the received signal may be a function of the channel transferfunction, or given by

${X_{p}( {k,n} )} = {{\sum\limits_{j = 0}^{J - 1}{\sum\limits_{m = 0}^{M - 1}{{H_{{c:j},k}( {m\;\delta\;\tau} )}\;{V_{jk}( {{n\;\delta\; t_{nyquist}} - {m\;\delta\;\tau}} )}\;\delta\;\tau}}} + {v_{k,n}.}}$

Step 716 executes by revising the signal model. The focus of this stepis on associating with the k₀ ^(th) receiver sub-aperture. Usingsimplified notation and dropping the indices associated with thesub-aperture of receiver 300 and the coherent processing interval, thesignal model may be rewritten to

$Y_{n} = {{\sum\limits_{j = 0}^{J - 1}{\sum\limits_{m = 0}^{M - 1}{H_{j,m}S_{j,{n - m}}}}} + {v_{n}\mspace{14mu}{where}}}$Y_(n) ≡ X_(p)(k, n) H_(j, m) ≡ H_(c : j, k)(m δ τ) δ τS_(j, n) ≡ V_(jk)(n δ τ_(nyquist))

Thus, the signal model for a batch of data associated with the k₀ ^(th)receiver sub-aperture and the time sample n=n₀ ma be given byY≡{X_(k) ₀ _(,n)}_(nε[n) ₀ _(,n) ₀ _(+N−1])

Step 718 executes by generating a linear system model for the signalmodel. The linear system model may be expressed as

$\begin{pmatrix}{Y(0)} \\{Y(1)} \\ \\{Y( {N - 1} )}\end{pmatrix} = {{\sum{\bullet\begin{pmatrix}H_{0,0} \\H_{0,1} \\ \\H_{0,{M - 1}} \\H_{1,0} \\H_{1,1} \\ \\H_{1,{M - 1}} \\{\;\underset{\_}{{\bullet\mspace{11mu}\bullet\mspace{11mu}\bullet}\mspace{31mu}}} \\H_{{J - 1},0} \\H_{{J - 1},1} \\ \\\underset{\_}{H_{{J - 1},{M - 1}}}\end{pmatrix}}} + \begin{pmatrix}{v(0)} \\{v(1)} \\ \\{v( {N - 1} )}\end{pmatrix}}$

The general least squares solution may be given by {tilde over(H)}=(Σ^(T)Σ)⁻¹Σ^(T)Y where P=Σ^(T)Y represents application of ageneralized matched filter to received data and R≡(Σ^(T)Σ) is awhitening, or waveform compensation filter, such as waveformcompensation filter 408. When N_(t)=J·M·N, Σ may be invertible and,thus, H=Σ⁻¹Y.

The disclosed process may remove the pseudo-random baseband modulationfrom the received signal. The channel transfer function may still haveRF phase delay information encoded within. The phased delay informationmay be related to the direction of the clutter patch relative to thespecific transmitter and receiver sub-apertures. The application of thesteering vectors to the channel transfer functions may separate out thelinear phase delays associated with specific directions from transmitter200 and receiver 300. Application of transmitter aperture weights to theBCTF compensates for the linear phase term in the BCTF model. Thisaction may provide a technique to resolve scattering sources in thedelay and the angle of the received signal.

FIG. 8 depicts a flowchart for performing radar operations within anadaptive broadcast radar system in accordance with an embodiment of thepresent invention. FIG. 8 depicts the processes disclosed above, withreference to overall adaptive broadcast radar system. Within the system,a signal waveform is transmitted continuously from a transmitter over anarea that may have a number of receivers. The receiver may receivesignals from the transmitter and scattered signals reflected off atarget within the area.

Step 802 executes by generating a signal waveform within thetransmitter, such as transmitter 200. The signal waveform may begenerated within a sub-aperture of transmitter 200. Preferably, eachsub-aperture of transmitter 200 may generate a signal in a pseudo-randommanner. Step 804 executes by encoding the signal waveform with numericalinformation. Specifically, the signal waveform is encoded orthogonally.By being encoded orthogonally, the signals from the differentsub-apertures may be distinguishable. Further, the degrees of freedomassociated with transmitter 200 also may be encoded into the signals.Step 806 executes by placing phase shifts and amplitude, or weights,onto the signal. The phase shifters and weights may be set independentlyfor each sub-aperture of transmitter 200.

Step 808 executes by transmitting the orthogonally encoded signal in acontinuous manner over an area. Preferably, the area may be forward oftransmitter 200. The signals may be directed towards the area with thepurpose of reaching receivers within the area. Further, transmitter 200transmits from an array antenna coupled to the sub-apertures.Transmitter 200 may be in motion, such as an airplane pointingtransmitter 200 below itself to receivers on the ground. Step 810executes by receiving the direct and scattered signals at receiver 300.The received signal may be a composite of the transmitted signals, andmay have data for the transmitted signals. Receiver 300 also may have anarray antenna coupled to sub-apertures that correlate to thesub-apertures on transmitter 200. The signals received are digitallyreconstructed to determine target parameters from the direct pathsignals and scattered signals.

Step 812 executes by performing motion compensation to remove timedependent phase delays between transmitter 200 and receiver 300. Motioncompensation may be performed independently for each sub-aperture ofreceiver 300. Step 814 executes by determining the delay and doppler, ifapplicable, for the received signal. This information may be placed intoa channel transfer function.

Step 818 executes by regenerating the transmit signal from transmitter200. By using the information encoded in the transmitted signalwaveforms, such as degrees of freedom and the transmit sub-aperture orelement number, and the information derived from the signal, such asdelay and doppler, the transmit beam may be recontructed. Although thetransmitted signal was not “beamed” to a specific location, receiver 300may use the parameters identified above to “reconstruct” a beamtransmitted at the target. The reconstructed beam may be used foradditional radar operations, such as target tracking. Step 820 executesby controlling the regenerated transmit beam at receiver 300 fortracking the target detected from the scattered signals.

It will be apparent to those skilled in the art that variousmodifications and variations can be made in the disclosed embodiments ofthe present invention without departing from the spirit or scope of theinvention. Thus, it is intended that the present invention embodies themodifications and variations of this invention provided that they comewithin the scope of the appended claims and their equivalents.

1. A method for obtaining target parameters within an adaptive broadcastradar system, comprising: coding information about a signal waveformgenerated by a transmitter having sub-apertures; receiving a signal at areceiver having sub-apertures corresponding to said sub-apertures ofsaid transmitter, wherein said received signal correlates to said signalwaveform; decoding information about said signal waveform from saidreceived signal; and determining a data quad from said decodedinformation, wherein said data quad includes degrees of freedomassociated with said transmitter.
 2. The method of claim 1, furthercomprising generating said signal waveform within said sub-aperture ofsaid transmitter.
 3. The method of claim 1, further comprising applyinga phase shift to said signal waveform within said transmittersub-apertures.
 4. The method of claim 3, further comprising applying aweight vector to said signal waveform within said transmittersub-apertures.
 5. The method of claim 4, further comprising motioncompensating said received signal by removing said weight vectors andsaid phase shifts.
 6. The method of claim 1, wherein said receivedsignal is a composite of transmitted signal from said signal waveform.7. The method of claim 1, further comprising generating a channeltransfer function comprising delay and doppler signal components of saidreceived signal.
 8. The method of claim 7, wherein said determiningincludes formatting said channel transfer function with a weight vectorand measurement covariance of said received signal.
 9. The method ofclaim 8, wherein said signal waveform is transmitted as an orthogonalwaveform.